Discrete audio amplifier basics - Part 2: JFETs, MOSFETs and other circuit
configurations

John Linsley Hood - March 24, 2010

[Part 1
covers the basics of bipolar junction transistors and looks at some typical BJT
audio circuit configurations.]

9.7 Junction Field Effect Transistors (JFETs)
JFETs are, almost
invariably, depletion mode devices, which means that there will be some drain
current at a zero-applied gate-source potential. This current will decrease in a fairly linear manner as the reverse
gate-source potential is increased, giving an operating characteristic, which
is, in the case of an N-channel JFET, very similar to that of a triode valve, as
shown in curve 'c' of Figure
9.3.

Like a thermionic valve, the operation of the device is limited to the range
between drain (or anode) current cutoff and gate (or grid) current. In the case
of the JFET, this is because the gate-channel junction is effectively a silicon
junction diode—normally operated under reverse bias conditions. If the gate
source voltage exceeds some 0.6 V in the forward direction, it will conduct,
which will prevent gate voltage control of the channel current.

P-channel JFETs are also made, although in a more limited range of types, and
these have what is virtually a mirror image of the characteristics of their
N-channel equivalents, although in this case the gate-source forward conduction
voltage will be of the order of -0.6 V, and drain current cutoff will occur in
the gate voltage range of +3 to +8 V. Although Sony did introduce a range of
junction FETs for power applications, these are no longer available, and typical
contemporary JFETs cover the voltage range (maximum) from 15 to 50 V, mainly
limited by the gate-drain reverse breakdown potential, and with permitted
dissipations in the range 250"400 mW. Typical values of gm
(usually called gfs in the case of JFETs) fall in the range of
2"6 mS.

JFETs mainly have good high-frequency characteristics, particularly the
N-channel types, of which there are some designs capable of use up to 500 MHz.
Modern types can also offer very low noise characteristics, although their very
high input impedance will lead to high values of thermal noise if their input
circuitry is also of high impedance; however, this is within the control of the
circuit designer. The internal noise resistance of a JFET, R(n), is
related to the gfs of the device by

R(n) ohms ≈ 0.67/gfs

and the value of gfs can be made higher by paralleling a
number of channels within the chip. The Hitachi 2SK389 dual matched-pair JFET
achieves a gfs value of 20 mS by this technique, with an
equivalent channel thermal noise resistance of 33 ohms.2

JFETs (cont.)
Although JFETs will work in most of the circuit
layouts shown for junction transistors, the most significant difference in the
circuit structure is due to their different biasing needs. In the case of a
depletion mode device it is possible to use a simple source bias arrangement,
similar to the cathode bias used with an indirectly heated valve, of the kind
shown in Figure 9.13.

Figure 9.13: Simple JFET biasing system.

As before, the source resistor, R3, will need to be
bypassed with a capacitor, C2, if the loss of stage gain due
to local NFB is to be avoided. As with a pentode valve, which the junction FET
greatly resembles in its operational characteristics, the simplest way of
calculating stage gain is by the relationship:

A ≈ gfsRL.

The device manufacturers will frequently modify the structure of the JFET to
linearize its Vg/Id characteristics, but, in
an ideal device, these will have a square-law relationship, as defined by

gfs = Id/Vg ≈ [1 -
(Vgs/Vgc)] ≈ /Vg.

For a typical JFET operating at 2 mA drain current, the gfs
value will be of the order of 1"4 mS, which would give a stage gain of up to 40
if R2, in Figure 9.13, is 10 kΩ. This is very much lower than
would be given by a BJT and is the main reason why they are not often used as
voltage-amplifying devices in audio systems unless their very high input
impedance (typical values are of the order of 1012 Ω) or their high,
and largely constant, drain impedance characteristics are advantageous.

The real value of the JFET emerges in its use with other devices, such as the
bipolar/FET cascode shown in Figure 9.14 or the FET/FET cascode layout of Figure
9.15.

Figure 9.14: Bipolar/FET cascode.

Figure 9.15: FET/FET cascode.

In the first of these, use of the JFET in the cascode connection confers the
very high output impedance of the JFET and the high degree of output/input
isolation characteristic of the cascode layout, coupled with the high stage gain
of the BJT. The source potential of the JFET (Q2) will be
determined by the reverse bias appearing across the source/gate junction, and
could typically be of the order of 2"5 V, which will define the collector
potential applied to Q1.

A further common application of this type of layout is that in which the
cascode FET (Q2 in Figure 9.15) is replaced by a high-voltage
BJT. The purpose of this arrangement is to allow a JFET amplifier stage to
operate at a much higher rail voltage than would be allowable to the FET on its
own; this layout is often found as the input stage of high-quality audio amps. A
feature that is very characteristic of the JFET is that for drain potentials
above about 3 V, the drain current for a given gate voltage is almost
independent of the drain voltage, as shown in Figure 9.16.

Figure 9.16: Drain current characteristics of junction
FET.

BJTs have a high characteristic collector impedance, but their
Ic/Vc curve for a fixed base voltage, also
shown, for comparison, in Figure 9.16, is not as flat as that of the JFET. The
very high dynamic impedance of the JFET resulting from this very flat
Id/Vd relationship encourages the use of
these devices as constant current sources, shown in Figure 9.17.

Figure 9.17: Current source layout.

In this form the JFET can be treated as a true two-terminal device, from
which the output current can be adjusted, with a suitable JFET, over the range
of several milliamperes down to a few microamperes by means of RV1.

MOSFETs
9.8 Insulated Gate FETs (MOSFETs)
Insulated gate
FET devices, usually called MOSFETs, are by far the most widely available, and
most widely used, of all the field effect transistors. They normally have a
rather worse noise figure than an equivalent JFET, but, on the plus side, they
have rather more closely controlled operating characteristics.

The range of types available covers the very small-signal, low-working
voltage components used for VHF amplification in TVs and FM tuners (for which
applications a depletion-mode dual-gate device has been introduced that has very
similar characteristics to those of an RF pentode valve) to high-power,
high-working voltage devices for use in the output stages of audio amplifiers,
as well as many other high-power and industrial applications. They are made in
both depletion- and enhancement-mode forms (the former having gate
characteristics similar to that of the JFET, whereas the latter description
refers to the style of device in which there is normally no drain current in the
absence of any forward gate bias), in N-channel and P-channel versions, and, at
the present time, in voltage and dissipation ratings of up to 1000 V and 600 W,
respectively.

All MOSFETs operate in the same manner, in which a conducting electrode (the
gate) situated in proximity to an undoped layer of very high purity
single-crystal silicon (the channel), but separated from it by a very thin
insulating layer, is caused to induce an electrostatic charge in the channel,
which will take the form of a layer of mobile electrons or holes. In
small-signal devices this channel is formed on the surface of the chip between
two relatively heavily doped regions, which will act, respectively, as the
source and the drain of the FET, while the conducting electrode will act as the
current controlling gate.

Although modern photolithographic techniques are capable of generating
exceedingly precise diffusion patterns, the length of the channel formed by
surface-masking techniques in a lateral MOSFET will be too long to allow a low
channel "on" resistance. For high current applications, the semiconductor
manufacturers have therefore evolved a range of vertical MOSFETs. In these, very
short channel lengths are achieved by sequential diffusion processes from the
surface, which are then followed by etching a V- or U-shaped trough inward from
the surface so that the active channel is formed across the exposed edge of a
thin diffused region. Because this channel is short in length, its resistance
will be low, and because the manufacturers generally adopt device structures
that allow a multiplicity of channels to be connected electrically in parallel,
channel "on" resistances as low as 0.008 Ω have been achieved.

Like a JFET, the MOSFET would, left to itself, have a square-law relationship
between gate voltage and drain current. However, in practice, this is affected
by the device geometry, and many modern devices have a quite linear
Id/Vg characteristic, as shown in Figure
9.18 for an IRF520 power MOSFET.

Figure 9.18: Power MOSFET.

The basic problem with the MOSFET is that of gate/channel overvoltage
breakdown, in which the thin insulating layer of silicon oxide or silicon
nitride between the gate electrode and the channel breaks down. If this happens
the gate voltage will no longer control the drain current and the device is
defunct.

Because it is theoretically possible for an inadvertent electrostatic charge,
such as might arise with respect to the ground if a user were to wear nylon or
polyester fabric clothing and well-insulated shoes, it is common practice in the
case of small-signal MOSFETs for protective diodes to be formed on the chip at
the time of manufacture. These could be either zener diodes or simple junction
diodes connected between the gate and the source or the source and drain, as
shown in Figure 9.19.

Figure 9.19: Diode gate protection.

In power MOSFETs, these protective devices are seldom incorporated into the
chip. There are two reasons for this: (1) that the effective gate/channel area
is so large that the associated capacitance is high, which would then require a
relatively large inadvertently applied static charge to generate a destructive
gate/channel voltage (typically >40 V), and (2) that such protective diodes
could, if they were triggered into conduction, cause the MOSFET to act as a
four-layer thyristor and become an effective electrical short circuit. However,
there are usually no performance penalties that will be incurred by connecting
some external protective zener diode in the circuit to prevent the gate/source
or gate/drain voltage exceeding some safe value; this is a common feature in the
output stages of audio power amplifiers using MOSFETs.

Apart from the possibility of gate breakdown, which, in power MOSFETs, always
occurs at less than the maker's quoted voltage, except at zero drain current,
MOSFETs are quite robust devices, and the safe operating area rating (SOAR)
curve of these devices, shown for a typical MOSFET in Figure 9.20, is free from
the threat of secondary breakdown whose limits are shown, for a power BJT, in Figure
9.12. The reason for this freedom from localized thermal breakdown in the
MOSFET is that the mobility of the electrons (or holes) in the channel decreases
as the temperature increases, which gives all FETs a positive temperature
coefficient of channel resistance.

Figure 9.20: Power MOSFET SOAR limits.

Although it is possible to propose a mathematical relationship between gate
voltage and drain current, with MOSFETs as was done in the case of the JFET, the
manufacturers tend to manipulate the diffusion pattern and construction of the
device to linearize its operation, which leads to the type of performance
(quoted for an actual device) shown in Figure 9.21 . However, as a general rule,
the gfs of a MOSFET will increase with drain current, and a
forward transconductance (slope) of 10 S/A is quoted for an IRF140 at an ID
value of 15 A.

Figure 9.21: MOSFET characteristics.

Power BJTs vs Power MOSFETs as Amplifier Output Devices
9.9
Power BJTs vs Power MOSFETs as Amplifier Output Devices
Some rivalry
appears to have arisen between audio amplifier designers over the relative
merits of power BJTs, as compared with power MOSFETs. Predictably, this is a
mixture of advantages and drawbacks.

Because of the much more elaborate construction of the MOSFET, in which a
multiplicity of parallel connected conducting channels is fabricated to reduce
the conducting "on" resistance, the chip size is larger and the device is
several times more expensive both to produce and to buy. The excellent HF
characteristics of the MOSFET, especially the N-channel V and U MOS types, can
lead to unexpected forms of VHF instability, which can, in the hands of an
unwary amplifier designer, lead to the rapid destruction of the output devices.

However, this excellent HF performance, when handled properly, makes it much
easier to design power amplifiers with good gain and phase margins in the
feedback loop, where overall NFB is employed. In contrast, the relatively
sluggish and complex characteristics of the junction power transistor can lead
to difficulties in the design of feedback amplifiers with good stability
margins.

Also, as has been noted, the power MOSFET is intrinsically free from the
problem of secondary breakdown, and an amplifier based on these does not need
the protective circuitry that is essential in amplifiers with BJT output devices
if failure is to be avoided when they are used at high power levels with very
low impedance or reactive loads. The problem here is that the protective
circuitry may cut in during high-frequency signal level peaks during the normal
use of the amplifier, which can lead to audible clipping. (Incidentally, the
proponents of thermionic valve-based audio amplifiers have claimed that the
superior audible qualities of these, by comparison with transistor-based
designs, are due to the absence of any overload protection circuitry that could
cause premature clipping and to their generally more graceful behavior under
sporadic overload conditions).

A further benefit enjoyed by the MOSFET is that it is a majority carrier
device, which means that it is free from the hole-storage effects that can
impair the performance of power junction transistors and make them sluggish in
their turn-off characteristics at high collector current levels. However, on the
debit side, the slope of the Vg/Id curve of
the MOSFET is less steep than that of the Vb/Ic curve of the
BJT, which means that the output impedance of power MOSFETs used as source
followers is higher than that of an equivalent power BJT used as an emitter
follower.

Other things being equal, a greater amount of overall negative feedback
(i.e., a higher loop gain) must therefore be used to achieve the same low
amplifier output impedance with a power MOSFET design than would be needed with
a power BJT one. If a pair of push"pull output source/emitter followers is to be
used in a class AB output stage, more forward bias will be needed with the
MOSFET than with the BJT to achieve the optimum level of quiescent operational
current, and the discontinuity in the push"pull transfer characteristic will be
larger in size, although likely to introduce, in the amplifier output signal,
lower rather than higher order crossover harmonics.

9.10 U and D MOSFETs
I have, so far, lumped all power MOSFETs
together in considering their performance. However, there are, in practice, two
different and distinct categories of these, based on their constructional form,
and these are illustrated in Figure 9.22.

Figure 9.22: MOSFET design styles.

In the V or U MOS devices - these are just different names for what is
essentially the same system, depending on the profile of the etched slot - the
current flow, when the gate layer has been made sufficiently positive (in the
case of an N-channel device) to induce a mobile electron layer, will be
essentially vertical in direction, whereas in D-MOS or T-MOS construction the
current flow is T shaped from the source metallization pads across the exposed
face of the very lightly doped P region to the vertical N-/N+ drain sink.

Because it is easier to manufacture a very thin diffused layer (=short
channel) in the vertical sense than to control the lateral diffusion width, in
the case of a T-MOS device, by surface masking, the U-MOS devices are usually
much faster in response than the T-MOS versions, but the T-MOS equivalents are
more rugged and more readily available in complementary (N-channel/P-channel)
forms.

All power MOSFETs have a high input capacitance, typically in the range of
500"2500 pF, and because devices with a lower conducting resistance
(Rds/on) will have achieved this quality because of the
connection of a large number of channels in parallel, each of which will
contribute its own element of capacitance, it is understandable that these low
channel resistance types will have a larger input capacitance.

Also, in general, P-channel devices will have a somewhat larger input
capacitance than an N-channel one. The drain/gate capacitance - a factor that is
very important if the MOSFET is used as a voltage amplifier - is usually in the
range of 50"250 pF. The turn-on and turn-off times are about the same (in the
range 30"100 nS) for both N-channel and P-channel types, mainly determined by
the ease of applying or removing a charge from the gate electrode. If
gate-stopper resistors are used—helpful in avoiding UHF parasitic oscillation
and avoiding latch-up in audio amplifier output source followers—these will form
a simple low-pass filter in conjunction with the device input capacitance and
will slow down the operation of the MOSFET.

Figure 9.23: MOSFET symbols.

Although circuit designers tend to be rather lazy about using the proper
symbols for the components in the designs they have drawn, enhancement-mode and
depletion-mode MOSFETs should be differentiated in their symbol layout, as shown
in Figure 9.23. As a personal idiosyncrasy, I also prefer to invert the symbol
for P-channel field effect devices, as shown, to make this polarity distinction
more obvious.

Useful Circuit Components - Constant Current Sources & Current
Mirrors
9.11 Useful Circuit Components
By comparison with the
situation that existed at the time when most of the pioneering work was done on
valve-operated audio amplifiers, the design of solid-state amplifier systems has
been facilitated greatly by the existence of a number of circuit artifices,
contrived with solid-state components, which perform useful functions in the
design. This section shows a selection of the more common ones.

9.11.1 Constant Current Sources
A simple two-terminal
constant current (CC) source is shown in Figure
9.17, and devices of this kind are made as single ICs with specified output
currents. By comparison with the discrete JFET/resistor version, the IC will
usually have a higher dynamic impedance and a rather higher maximum working
voltage.

In power amplifier circuits it is more common to use discrete component CC
sources based on BJTs, as these are generally less expensive than JFETs and
provide higher working voltages. The most obvious of these layouts is that shown
in Figure 9.24(a), in which the transistor, Q1, is fed with a
fixed base voltage - in this case derived from a zener or avalanche diode,
although any suitable voltage source will serve - and the current through
Q1 is constrained by the value chosen for R1
in that if it grows too large, the base-emitter voltage of Q1
will diminish and Q1 output current will fall. Designers
seeking economy of components will frequently operate several current source
transistors and their associated emitter resistors (as
Q1/R1) from the same reference voltage
source.

Figure 9.24: Constant current sources.

In the somewhat preferable layout shown in Figure 9.24(b), a second
transistor, Q2, is used to monitor the voltage developed
across R1 due to the current through Q1;
when this exceeds the base emitter turn-on potential (about 0.6 V),
Q2 will conduct and will steal the base current to
Q1 provided from Vref through
R2.

In the very simple layout shown in Figure 9.24(c), advantage is taken of the
fact that the forward potential of a P-N junction diode, for any given junction
temperature, will depend on the current flow through it. This means that if the
base-emitter area and doping characteristics of Q1 are the
same as those for the P-N junction in D1 (which would,
obviously, be easy to arrange in the manufacture of ICs), then the current
(iout) through Q1 will be caused to mimic that flowing through
R1, which is labeled iref. This particular
action is called a current mirror, and several further versions of these are
shown in Figure 9.25.

Figure 9.25: Current mirror circuits.

9.11.2 Current Mirror Layouts
Current mirror (CM) layouts
allow, for example, the output currents from a long-tailed pair to be combined,
which increases the gain from this circuit. In the version shown in Figure
9.25(a), two matched transistors are connected with their bases in parallel so
that the current flow through Q2 will generate a base-emitter
voltage drop that will be precisely that which is needed to cause
Q1 to pass the same current.

If any doubt exists about the similarity of the characteristics of the two
transistors, as might reasonably be the case for randomly chosen devices, the
equality of the two currents can be assisted by the inclusion of equal value
emitter resistors ( R1,R2) as shown in
Figure 9.25(b). For the perfectionist, an improved three-transistor current
mirror layout is shown in Figure 9.25(c). Commonly used circuit symbols for
these devices are shown in Figures 9.25(d) and 9.25(e).

Circuit Oddments
9.12 Circuit Oddments
Several circuit
modules have found their way into amplifier circuit design, and some of the more
common of these are shown in Figure 9.26. Both the DC bootstrap, shown in Figure
9.26(a), and the JFET active load, shown in Figure 9.26(b), act to increase the
dynamic impedance of R1, although the DC bootstrap, which can,
of course, be constructed using complementary devices, has the advantage of
offering a low output impedance.

Figure 9.26: Circuit oddments.

The amplified diode, shown as Figure 9.26(c), is a device that is much used
as a means of generating the forward bias required for the transistors used in a
push" pull pair of output emitter followers, particularly if it is arranged so
that Q1 can sense the junction temperature of the output
transistors. It can also be used, over a range of relatively low voltages, as an
adjustable voltage source to complement the fixed voltage references provided by
zener and avalanche diodes, band-gap references (IC stabilizers designed to
provide extremely stable low voltage sources), and the wide range of voltage
stabilizer ICs.

Finally, when some form of impedance transformation is required, without the
Vbe offset of an emitter follower, this can be contrived as
shown by putting two complementary emitter followers in series. This layout will
also provide a measure of temperature compensation.

9.13 Slew Rate Limiting
This is a potential problem that can occur
in any voltage amplifier or other signal handling stage in which an element of
load capacitance (which could simply be circuit stray capacitance) is associated
with a drive circuit whose output current has a finite limit. The effect of this
is shown in Figure 9.27.

Figure 9.27: Cause of slew rate limiting.

If an input step waveform is applied to network (a), then the output signal
will have a waveform of the kind shown at 'a', and the slope of the curve will
reflect the potential difference that exists, at any given moment, between the
input and the output. Any other signal that is present at the same time will
pass through this network, from input to output, and only the high-frequency
components will be attenuated.

However, if the drive current is limited, the output waveform from circuit
9.27 (b) will be as shown at 'b' and the slope of the output ramp will be
determined only by the current limit imposed by the source and the value of the
load capacitance. This means that any other signal component that is present, at
the time the circuit is driven into slew rate limiting, will be lost. This
effect is noticeable, if it occurs, in any high-quality audio system and gives
rise to a somewhat blurred sound - a defect that can be lessened or removed if
the causes (such as too low a level of operating current for some amplifying
stage) are remedied.

It is prudent, therefore, for the amplifier designer to establish the
possible voltage slew rates for the various stages in any new design and then to
ensure that the amplifier does not receive any input signal that requires rates
of change greater than the level that can be handled. A simple input integrating
network of the kind shown in Figure 9.27(a) will often suffice.

Printed with permission from Newnes. A division of Elsevier. Copyright 2009.
"Audio
Engineering: Know It All" by Douglas Self et al. For more information about
this book and other similar titles, please visit http://www.newnespress.com/.

References:
1. Linsley Hood, J., Wireless World, 437-441,
September, 1971.
2. Linsley Hood, J., 'Low noise systems', Electronics Today
International, 42-46, 1992.

Related links:
Discrete
audio amplifier basics - Part 1: Bipolar junction transistor circuits
Distortion
in power amplifiers, Part II: the input stage | Part
III: the voltage-amplifier stage
Analog
Circuits: World Class Design, Part 2 | Part
3
MOSFET
or bipolar, which should you use?