The
Emperor's New Amplifier
TM

by
Norman L. Koren

Updated June 4,
2003

Norman Koren Vacuum tube
audio page | Photography
page
Feedback
and Fidelity | Improved
vacuum tube models for SPICE simulations

November 3,
2002: I've added diagrams
of the hand-wired circuit boards and more footnotes to the main
schematic.
February 2003: Paul
Heggeseth is
building a version of TENA (brave soul). Check his site
for updates.

Table of
Contents

Features
| Schematic
| Design
philosophy | Input
stage/phase inverter
Toroidal
output transformer | Bias
servo and adjustment | Feedback
| Class AB2
output and drivers
Power
supplies | MOSFET
regulators | Minimum
feedback modification | Performance

Sound
| Additional
notes: underside and circuit board diagrams.
PSpice
output

I spent much of 1997 designing and
building my ultimate dream amplifier, which I named "The Emperor's New
Amplifier"TM (TENA) for a quality it shares with the fabled
emperor's wardrobe-- transparency. It was also an oblique reference to the
marketing hype that pollutes high-end audio. I thought about commercializing
it, especially when the lab where I worked announced it was shutting down,
but I soon realized that marketing high-end tube amplifiers is not a reliable
way to make a living. (I'm paycheck-addicted.) Counterpoint
Electronics, a high-end audio manufacturer located in a 45,000 square
foot building five miles from where I lived at the time, vanished overnight.
Then I thought I might write a magazine article, but the move to Colorado
distracted me.

TENA has been working reliably since 1997, and
in a way it's ruined me as an audiophile. I've become contented. I'm so
pleased with it I've abandoned the quest for audio perfection. (Photography
keeps me busy enough.) Now the time has come to share it-- via the Internet.
Brace yourself for a tour de force of amplifier design. It's arguably
the most sophisticated vacuum tube amplifier ever. But
be forewarned if you plan to build it: It's not a simple project. Absolutely
not for beginners!

Features

Output: 4-6550's in triode-mode class AB2
push-pull parallel. About 80 watts RMS per channel.

No global negative feedback. Several local
loops with limited negative feedback.

Ultra-wide bandwidth Plitron
toroidal output transformer.

Servo to maintain precise dc-balance in
the output circuit.

MOSFET-regulated power supplies.
Relative rather than absolute voltage reference.

Designed using extensive PSpice computer
simulation.

Constructed as a pair of
monoblocks.

Schematic
diagram
If the schematic diagram is too large to print
directly from your Web browser, save it as a GIF file and print it on an image
editor or viewer. If you'd like to save a larger and more detailed version of
the schematic, shift-click on TENA_Sch_1BIG.gif.

TENA schematic diagram (simplified power supply)

Notes on the schematic diagram

The power supply has been simplified-- Power transformers and rectifiers
have been omitted and some parts have been omitted from the MOSFET voltage
regulator circuits: 1N5242 zener diodes between the source and gate and 10k
resistors in series with the gate. These parts serve as protection in case of
accidental short circuits, but don't affect the operating point. The full
power supply schematic is shown below.

6SN7's are used instead of 12AU7's for the driver tubes. They have the
same plate characteristics, but they have higher maximum plate voltage (450
vs. 330 V) and greater plate dissipation (3.75 vs. 2.75 W per section). Think
of the octal-based 6SN7 as a 12AU7 on steroids.

The NODESET blocks (lower right) initialize dc levels in the bias servo so
the simulation runs properly. They don't exist as physical entities.

All resistors are 1/2 watt unless noted, and with the following
exceptions. R3P, R4P and R5C through R8C (68K) are 2W. R9C through R12C (20
ohm) are 1W: I used 2-10 ohm resistors in series to make them. R9S through
R12S (200 ohms) are 2W.

Capacitor voltage ratings: It never hurts to go over the minimum, though
the capacitors will be larger and may cost more. I often use caps with higher
voltage ratings because I have them on hand or found them at a good price in
an electronics surplus shop. Here are some minimum ratings: C1G: 100V. C2G:
400V (600V would be better; necessary without the time delay); C3G and C4G:
400V (600V would be better); C3M and C4M: 400V; C5G through C8G: 600V; CBS2,
CBS4, CBS6 and CBX1 through CBX3: 100V. The voltage ratings of many of the
power supply capacitors are shown on the circuit board wiring diagrams, below.

Some of the feedback connections may be a little hard to trace. ORN goes
between the 20 ohm output transformer secondary feedback winding and R3F near
TU3. Similarly, VLT goes to R4F near TU4. BLU and BLK on the output
transformer secondary speaker winding go to the output tube cathode
circuits.

RLS (5 ohms) is a simulated speaker load.

CRF: All three nodes are connected together.

BRN and VIO are not used. They are the ultra-linear taps. In the original
version of TENA there was a switch to select output tube screen grid
connections between BRN and VIO (UL mode) and GRN and YEL (triode
mode).

Design
philosophy
In the 1990's I strove to learn the secrets of
designing circuits with superior audio quality. I'm technically curious and also
cheap-- I didn't want to spend big bucks on cosmetically impressive equipment of
dubious design. SPICE computer modeling became my primary learning tool because
it enabled me to observe inner details of amplifier operation not obvious from
outer measurements.

My goal was to design a high-powered wide-bandwidth amplifier with the finest
sound and to keep it simple-- but not too simple. I rejected single-ended (SE)
designs because of their low power and limited bandwidth. I wanted to see if I
could get similar sound quality-- perhaps better-- with an efficient push-pull
design. And I wanted to do it my way.

I aimed for wide bandwidth, but not too wide. Very low frequency
1/f noise and radio stations don't enhance music. Most of the coupling
circuits have an RC time constant around 0.05 seconds, equivalent to a -3 dB
frequency of 3 Hz-- low but not too low.

No net ac current is drawn from any power supply tap. Pairs of tubes draw
equal but opposite current. This effectively removes the power supply from the
signal path and reduces the need for voltage regulation. Nevertheless robust
MOSFET regulators are employed for ultimate sonic refinement.

I avoid global negative feedback-- a loop from the amplifier output to
its input. There are several local loops-- around one or at most two gain
stages. Local feedback loops have numerous advantages and no adverse effect on
sound quality. Stability is much easier to maintain. The extreme bandwidth of
the output transformer allowed me to put a moderate amount of feedback in the
output circuit while maintaining excellent bandwidth and stability.

I originally designed TENA to be switchable between triode mode (plate
connected to screen grid) and ultra-linear (UL) mode (plate connected to output
transformer 40% taps). Output modes are discussed in Feedback
and Fidelity, Part 2. In conventional class AB1 operation, UL mode has much
greater output power than triode mode, but it has higher output impedance and
poorer linearity. In class AB2 operation (where grid current is drawn when the
grid is driven positive with respect to the cathode) I got plenty of power in
triode mode, so I eliminated the switch. (Duncan's
Amp Pages has a nice little discussion of operating classes.) A triode
behaves like a pentode with built-in local negative feedback. A
pentode's screen grid suppresses feedback from the plate, but it can be
restored by connecting it in triode mode (connecting the screen grid to the
plate instead of a fixed high voltage). Many audiophiles despise negative
feedback (see Feedback and
Fidelity), but ask them which tube type they prefer...

Try as I might I couldn't keep it simple. Class AB2 operation requires
individual cathode follower drivers, which requires extra power supplies. So it
got complex, but every part has a purpose; none is superfluous. I did nothing
because conventional wisdom told me to.

Components are operated conservatively-- well under maximum power dissipation
and voltage ratings. This ensures maximum reliability.

I tried to avoid exotic, hard to get, or expensive parts. I chose the 6550C
over "real" triodes because it's moderately priced, easy to find, rugged and has
high power dissipation. It behaves like a real triode when connected in triode
mode. Most "real" triodes have directly heated filaments which are difficult to
integrate into feedback circuits.

Several aspects of the design philosophy implemented in TENA-- maximizing
stability, minimizing
RF interference (which can result in"gritty" sound and "listener fatigue"),
and soft
clipping-- are discussed in detail in Feedback
and Fidelity. In writing this article I've become aware of a few things I
may change if I have the time or make TENA into a commercial product, but I have
no such plans at the moment.

Input stage/phase inverter
Input stage TU1
is a simple voltage gain stage with local negative feedback, derived from the
R1B, R1C voltage divider. It is capacitively coupled to split load phase
inverter TU2. The capacitor has an unusually low value-- 0.01 µF-- because TU2
has an exceptionally high input impedance-- several Megohms. The advantage of
capacitive coupling is that it allows the voltage level in TU2 to be set for
maximum output and it allows the ac current in TU2 to be precisely equal to, but
180 degrees out of phase with, the current in TU1. The net ac current drawn by
these two tubes from V+420 is therefore zero. This is an effective way of
isolating the audio signal from the power supply, which doesn't need to supply
ac current. In conventional designs ac signal often has to flow through
electrolytic capacitors, which are grungy leaky devices with memory-- harmful to
audio quality. I designed TENA to draw zero net ac current from all power supply
outputs (easy to do in a push-pull design), at least up to the power level
where one of the output tube pairs starts cutting off.

Toroidal output transformer
I chose the Plitron toroidal transformer because of its
exceptional bandwidth: -3 dB at over 200 kHz, the result of high primary
inductance (the good stuff) and low leakage inductance (the bad stuff-- kind of
like HDL and LDL cholesterol)-- much better than can be achieved with a
conventional EI transformer. High bandwidth is important because output
transformers have an intrinsic second order rolloff, which can make them
unstable in the presence of negative feedback unless careful phase compensation
is applied (see Feedback
and Fidelity). Phase compensation reduces the bandwidth, which is not a
problem with the Plitron toroidal transformers. But this bandwidth comes at a
price-- toroidal transformers are much less tolerant of dc-imbalance than EI
transformers; they may saturate at dc imbalances as low as 8 mA. (I don't know
the exact number; I never simulated it.) You would have to set the bias of each
tube individually, and then you'd have to worry about how the tubes age. So I
designed a bias servo circuit to maintain perfect dc-balance under all
circumstances except outright tube failure.

The Plitron PAT 4006CFB 100 Watt toroidal output transformer is not currently
listed on Plitron's website, but I've heard (June 2003) that it is available.
Contact Norman Woo. The closest models are
the 4006, which lacks the special feedback winding, and the 2100-CFB which has a
higher primary impedance. The minimum feedback version of TENA (below)
works with the 4006.

I won't go into detail about modeling toroidal transformers-- Plitron has a
treasure
trove of fine articles on its Website, and I said a few things in Improved
SPICE models.

Bias servo and adjustment
The
time-averaged (low pass filtered) dc current of an output tube operating in
class AB fixed bias is relatively constant at low power levels but increases at
high power levels. For this reason a fixed voltage cannot be used as a reference
for biasing the output tubes. One tube (TU9, driven by TU5) operates at fixed
bias, and its low pass filtered cathode voltage (CRF) is used as the reference
for biasing the other tubes.

The bias servo is illustrated in the lower left of the schematic. It uses the
LM324 quad op amp-- cheap but perfectly adequate. Inputs U1A, U1B and U1C of the
LM324 compare cathode voltages 10C, 11C, and 12C with reference voltage CRF,
which is the voltage on cathode 9C low pass filtered with RBS2 = 33k and CBS1 =
10µF ( located near U1B on the schematic). The LM324 outputs control the
P-channel MOSFETs, each of which controls a voltage divider between VBB (-90V)
and VOP (+12.5V) to deliver the appropriate bias voltage to the driver grid
circuits (BIAS_6, BIAS_7, and BIAS_8). This measures between -45 and -50V in my
amplifiers, which operate at 60 mA plate current. Audio purists please note: the
servo operates at extremely low frequencies; the op amp and MOSFETs are well
outside the audio signal path.

A single potentiometer, RB5 (in the VBB supply, bottom center), controls the
bias current directly in TU9, and all the other tubes indirectly through the
servo. Bias current may be measured across any of the 20 ohm resistors R9C-R12C
as E/20. They should all be the same if the servo is working properly. 1
to 1.2 volts is a good nominal value, corresponding to 50 to 60 mA per tube (70
mA was used in the Dynaco Mark III). Increasing the current increases power
consumption and reduces tube life and output power, but moves you closer to
Class A (where both tubes always conduct).

Feedback
The main
feedback loop goes from the 20 ohm cathode feedback winding on the output
transformer secondary (ORN, VLT) to the cathode circuits of the second gain
stages, TU3 and TU4. This is a much shorter path than a typical feedback loop,
which returns to the first input stage, and it is also balanced. Resistors R3C
and R3F on the cathode of TU3 and R4C and R4F on the cathode of TU4 form a
voltage divider than determines the amount of feedback-- about 12 dB for the
output stage in triode mode. The feedback loop is stabilized by 15pF Miller
capacitors C3M and C4M. This is a much better approach than the common technique
of putting the stabilization capacitor(s) in parallel with the feedback
resistor(s), which feeds RF signals picked up by the speaker wires back into the
input stages. This issue is discussed extensively in Feedback
and Fidelity. Output is virtually unchanged with 1µF is placed in parallel
with the 5 ohm loudspeaker RLS.

The cathodes (including the feedback winding center tap
WHT) are referenced to VBB = -90V. This gives TU3 and TU4 the huge voltage swing
required for the zero-gain cathode followers that drive the triode output stage.
The large voltage swing is the reason that the 6SN7 was chosen over the 12AU7,
which has a similar plate curve. If the output stage were operated in
ultra-linear mode, it would have higher gain and wouldn't require as large a
voltage swing on TU3 and TU4.

There is also a lesser feedback loop between the
speaker windings (BLU, BLK) and the output tube cathodes. This loop provides
less than 2 dB of feedback-- a very small amount. If I were an
anti-feedback purist, I'd eliminate the main loop (to the cathodes of TU3 and
TU4) as described below, but I'd keep
this one, which qualifies as a local feedback loop because it involves only one
amplification stage-- the output stage.

Class AB2 output stage and drivers
Class
AB2 differs from the more common class AB1 in that the output stage grid is
driven positive-- it draws grid current-- at high power levels. Class AB2 has no
advantage for output tubes operating in pentode mode and little advantage for
ultra-linear mode. But it results in a huge power boost for output tubes
operating in triode mode. You can get almost as much power out of class AB2
triodes as you can out of class AB1 pentodes.

If you try to do operate in class AB2 with conventional capacitive coupling,
the coupling capacitor starts charging as soon as grid current is drawn. This
drives the grid negative-- toward cutoff, and it recovers with the RC time
constant of the coupling capacitor and grid resistor. To operate successfully in
class AB2, the output stage must be either transformer or direct coupled. I
chose direct coupling because interstage transformers are expensive and have
limited bandwidth.

The direct coupled drivers are the source of much of TENA's complexity.
Because the quiescent grid voltage of each output tube must be set individually
to control its quiescent (dc) current, one driver tube (TU5-TU8) is required
for each output tube (TU9-TU12). Cathode followers (CF's) were chosen because
they have low output impedance and can source the needed output tube grid
current. The cathodes have to be somewhere near -50V to properly bias the
output tubes. This means the CF must be driven by voltages outside the range of
conventional power supplies, hence the need for VDR- and VDR+: the price of
perfection. In reviewing the design I find that the driver tubes may be
operating a little too conservatively-- dissipating only 0.78 W (of a
6SN7 maximum of 3.75 W). I've discussed driver dissipation under PSpice
output, below. I may increase VDR+ from 205 to around 250 V by increasing
RD1 from 470k to 680k. This would reduce the power dissipation in MOSFET MD1.

Output tube grid stop resistors R9G-R12G play an important role in TENA's
soft clipping. When power levels become high enough level for grid current to
be drawn, a voltage drop across these resistors gradually limits the plate
current. Soft clipping consists of low order harmonics which have much less
adverse effect on sound quality that the high order harmonics characteristic of
abrupt clipping. But total harmonic distortion for soft clipping amplifiers
tends to be higher. Yes, lower harmonic distortion doesn't mean better sound.
See "The great harmonic distortion scam" in Feedback
and Fidelity. TENA oscillated when the grid stop resistors were removed.
This was the only performance feature PSpice didn't catch. The reason is
that the output transformer model is somewhat simplified-- it's extremely
difficult to model its distributed capacitance.

Power supplies

Power supply detail

The time delay circuit (U3 (the 555B chip), Q1, Relay_SPDT_nb, RT1, CT1, CT2, RT3, D1, RV1, and
RT4)  has apparently never been implemented. RT4 should be replaced by a
straight wire; VBIN is connected directly to NTC (negative temperature
coefficient; 50 ohms cold; Mouser527-3504-50)
thermistor RV10.

The precise values of most of the capacitors in the
power supply, particularly CV1, CV2, CB1, CB2, CD1 and CD2, are not critical. In
many cases they were determined by parts availability. If the values are 2 uF or
under they are film capacitors. If they are over 2 uF they are
electrolytics.

Depending how you count there are two (power
transformers), four (rectifier circuits) or six (voltage levels). All use fast
recovery rectifier diodes. All except VDR- are taken from the mighty Plitron
854710 toroidal power transformer, which I can't seem to find in their catalog.
Toroidal power transformers perform well, but they have less of an advantage
than toroidal output transformers-- you don't need wide bandwidth for 60 Hz.
The CL80 inrush current limiter limits turn-on current in the tube
filaments.

NAME
Location

POWER
SUPPLY DESCRIPTION

V+450
Top

Main high voltage supply for output tubes
(TU9-TU12). About 450V unregulated. Uses 4 MUR4100's in a bridge. NTC
thermistor RV10 (Mouser527-3504-50)
limits relay turn-on current. Uses two large filter capacitors CV1 and
CV2 and choke (inductor) LV1. V+450 is the input to V+420V and VDR+.
[ A circuit employing a 555 timer was
designed to delay turn-on for about 20 seconds, giving tube cathodes time
to warm up. The intent was to prevent a turn-on voltage surge which could
potentially damage components such as filter and coupling capacitors. RT4
allows the supply voltage to gradually build up before the relay turns on.
This protects the relay contacts by reducing the transient turn-on
current. Its value (15k) was not highly optimized. In November 2002 I
examined TENA, and found that the time delay circuit had never been
implemented. Reliably is fine without it. Amazing what I've forgotten
since 1997. ]

V+420
Bottom
center

420V regulated supply for input stages
(TU1-TU4). Could be considered part of the main high voltage supply.
Regulated by IRF820 N-channel MOSFET with a modest heat sink.

VDR+
Bottom
right

Positive supply for driver tubes (TU5-TU8).
About 200V. The IRF820 N-channel MOSFET regulator is controlled by a
voltage divider (RD1, RV6) in the 420V supply. Requires a substantial heat
sink. In reviewing the design I see it might be slightly perferable to to
source VDR+ from V+420.

VDR-
Center
left

Negative supply for driver tubes (TU5-TU8).
About -305V.  Uses a separate isolation transformer (1:1 voltage
ratio) with a voltage doubler. Unregulated.

VBB
Center

-90V negative bias supply for output tubes, also
connected to TU3 and TU4. Uses voltage doubler taken from power
transformer bias winding. Regulated by IRF9610 P-channel MOSFET with a
modest heat sink.

VOP
Center
right

12.5V regulated supply for bias servo. Uses
voltage doubler taken from filament winding and LM317T
regulator.

I tried and failed to keep it simple, but the
sophisticated power supply is one of the keys to TENA's superb sound
quality.

MOSFET regulators
Thanks to Dennis Lusis for teaching me this technique. A MOSFET
(Metal Oxide Semiconductor Field Effect Transistor) is a three element device
whose terminals are a source, a drain, and an infinite impedance gate, shown
respectively on the bottom, top, and left of the MOSFETs in the schematic
diagram. You may think of the common N-channel MOSFET (IRF820 in TENA) as a
valve that maintains the source at a constant voltage (between 1 and 2 volts)
below the gate as long as the drain is several volts higher. The less common
P-channel MOSFET (IRF9610 in TENA) has the opposite polarity-- the source is
maintained at a voltage above the gate. At high frequencies this picture
is mucked up by gate capacitance.

The 420V supply (V+420) on the bottom of the power
supply schematic is a good example. Output voltage (V+420) is controlled by the
gate of MV1-- set by the RV5, RV6+RD1 voltage divider and the RV7, CV4 low pass
filter. RV7 can have a very high resistance (2.2 Megohms) because the gate has
infinite impedance. This allows the use of a film capacitor instead of a leaky,
noisy, unreliable electrolytic for CV4. When I hooked TENA to a pair of
extremely efficient loudspeakers (normally driven by SE triodes), I heard a very
slight hum. Because of this I would consider redesigning the low pass filter in
two stages (adding an R and C) for a future iteration of TENA.

The primary function of voltage regulators is to
maintain constant supply voltage independent load current. They are particularly
useful when a large voltage drop is required, as in the VDR+ supply, which drops
over 400V to 200V. TENA uses relative rather than absolute
regulation. The reference for a relative regulator is a multiple of the line
voltage. It it's done well, the reference voltage will change slowly and without
transient noise in response to changes in the line voltage. An absolute
regulator requires a fixed voltage reference. High voltage fixed references,
suitable for vacuum tube circuits, have their share of problems. For example
zener diodes can be plagued by noise and temperature coefficient issues. Since
the amplifier's gain is hardly affected by the supply voltage, I prefer
relative regulation; I know of no disadvantages. It's simple, reliable
and quiet. Some other MOSFET regulator designs can be found in Duncan's
Amp Pages and the Mods section of Welborne
Labs' catalog.

Minimum feedback modification
Since triodes have low
enough output impedance to drive most loudspeakers without negative feedback,
it's tempting to look at how TENA would perform with the main negative feedback
removed-- without the feedback from the secondary cathode feedback widing (ORN,
VLT) to the cathode circuits of TU3 and TU4. Feedback can interact with
difficult loads in nasty ways. Even though TENA has been designed for extreme
stability-- response is virtually unchanged with 1 µF in shunt with the
loudspeaker-- I haven't looked at every possible difficult load, like
inductive loads. And of course many audiophiles dislike the audible effects of
feedback (perhaps because it's often implemented badly).

Removing this loop would increase TENA's gain a bit
much, so I apply local feedback (the good kind with no adverse side effects) to
the cathode circuits of TU3 and TU4. To eliminate the main feedback loop, remove
all conections from the output transformer secondary cathode feedback winding
(ORN, VLT) and remove C3M and C4M. Reconfigure R3C and R3F as shown on the
right. Changes to TU4 should exactly mirror the changes to TU3.

I haven't tried this yet. I'll report on it if and when
I do. Simulated frequency response is virtually unchanged. One significant
advantage: You can use the Plitron
PAT 4006 output transformer, which is a part of their current product
line.

Simulated
performance
Actual measurements are
very close to the simulations, but simulated measurements are much easier to
display on a Web page.

PSpice simulated TENA frequency response in dB at TU1 plate (1P;
green),
TU2 plate and cathode (2P and 2C; red, blue), and Speaker
(SPKR; yellow)

Few tube amplifiers come close and none get better-- 1 dB down (±0.5 dB) at
the output (SPKR) at 9 Hz and 80 kHz with no irregularities at any of the
intermediate stages. Perfect match between phase inverter outputs below 10 kHz.
No more than 0.2 dB difference above 20 kHz. Response is stable even under
difficult loads. A 10 kHz square wave has only a slight overshoot and no ringing
when TENA is loaded with a 2 uF capacitor (a similar load to a large
electrostatic speaker) in parallel with a 5 ohm resistor. Most competitive
"high-end" amplifiers show severe ringing under these conditions.

PSpice simulated TENA time-domain response at driver grid (divided by 5;
red) and output
(SPKR; green), driven hard enough to illustrate soft
clipping, which starts at V(SPKR) = 24V,
equivalent to 116 Watts Peak
= 58 Watts RMS into the 5 ohm speaker load.

The PSpice model for soft clipping is rather crude; actual onset is more
gradual than the simulation indicates. Below 57 Watts RMS, distortion is
vanishingly low. It increases gradually above 57 Watts, but is of very low
order. That's why I tell people the output is "about" 80 Watts RMS. The
advantage of a design with gradually increasing low order distortion, as opposed
to an extremely linear design that clips abruptly, is discussed in Feedback
and Fidelity.

Output impedance is around 0.9 ohms. The power output and damping factor are
sufficient to drive nearly any loudspeaker.

Sound quality
I've read enough hyperbole
about amplifier sound quality to sink the Titantic without the benefit of an
iceberg. So I'll simply say it sounds wonderful-- extremely clear and dynamic.
It's everything I expected, though it's not as "sweet" as single-ended
amplifiers, which benefit from rather large amounts of second harmonic
distortion. The quest is over. I have other windmills to tilt (photography).
Fair Dulcinea awaits.

Norman
Koren Vacuum tube audio page | Photography
page
Feedback
and Fidelity | Improved
vacuum tube models for SPICE simulations

Additional notes
I
used point-to-point wiring on several perfboards: neither beautiful nor
manufacturable, but TENA has worked flawlessly since 1997. I would have designed
a PC board (actually, several) if I'd had the time to learn how. They're the
only way to go for manufacturability and they're also prettier, which is vital
for marketability. I used polypropylene capacitors throughout the audio path. I
didn't experiment with exotic capacitors or resistors. If you're interested in
the sonic effects of capacitors, check out Bob Pease's article on "Soakage"
in capacitors. The chassis was from Mark V Electronics in Los Angeles, now
sadly out of business. Size is 2 3/4x19"
(approx. 7x48 cm) for the front panel and
2.5x17x12" (approx. 6.3x43x30.5 cm)
for the main chassis.

TENA has three hand-wired perfboard circuit boards (Radio Shack 276-1395). PC
boards would be preferable, but the perfboards have been reliable. The scales
are in inches. The drawings were done in Corel Draw, which probably isn't ideal,
but it was on my computer and I knew how to use it. The illustrations show the
component side of the boards. The wiring on the opposite side is shown as thick
gray lines. When I built thems I printed out these diagrams along with mirror
images so I could see the wiring side in the proper orientation. The ruler shows
inches. GND is ground. The original Corel
draw files can be downloaded by shift-clicking on TENAboards.zip.

BIAS SERVO
BOARD  The wiring side is visible on the left in the
photograph, above. The socket is for the LM324 op amp. 3 IRF9610 MOSFETs are
used (the GDS rectangles). POT is RB5. The zener diodes (Znr), not shown in the schematic diagram above,
protect the MOSFETs and IC against static electricity events. The type is not
critical. The 1N5242 works fine. They were omitted because they have no effect
on the normal operation of the circuit; they would only complicate the
simulations.

POWER SUPPLY VOP,
VBB, TIME DELAY BOARD  The component side is visible
on the right in the photograph, above. GRN
and VIO are the bias secondary of the power
transformer, about 50V RMS. BLU and GRY are the heater windings of the power
transformer secondary. (I think the schematic should have shown 6.3V instead of
6.7V.) The 12.5V pin is VOP.
RLY_B goes to the relay that delays the
turn-on of the high voltage supply. I apparently changed RB1 from 470k (in the
schematic) to 330k. An LM317T voltage regulator (AOI) and an IRF9610 MOSFET (GDS),
both on small heat sinks, Mouser532-504102B00,
is used. The socket is for the 555B timer, which has been
omitted in the version of TENA I built, along with its associated circuitry
(parts close to the RLY_B pin: 1N4002, Q1
(EBC), 2.2M, 1K, 10UF, .01).

POWER SUPPLY +420V,
+250V BOARD  The wiring side (mostly) is visible near
the middle of the photograph, above. 2 IRF820 MOSFETs, mounted on heat sinks (a
large one, Mouser532-551002B00,
for MD1, on the left, which must drop a large voltage), are used. 470K
resistors RV6, RD1 and RD4 are rated at 1W.

PSpice output
This portion of the PSpice
ASCII output file shows voltages at key nodes. By knowing resistances between
nodes, current can be calculated from Ohm's law. For example, the cathode
resistor for one of the driver tubes, R5C = 68K, is connected between VDR- and
5C. The voltage drop across the resistor is -49.68-(-257.36) = 207.68 V. The
current is I = V/R = 3.05 mA. The power dissipation in this resistor is P = VI =
V2/R = I2R = 0.634 Watts. So I use a at least a 1 watt
resistor, with 2 watts recommended. The voltage drop across the 6SN7 tube is
VDR+ - V(5C) = 204.9+49.7 = 254.6 V. The power dissipated by the 6SN7 is VI =
254.6*0.00305 = 0.78 watts. The 6SN7 is operating extremely
conservatively: Maximum plate dissipation per plate with both operating
simultaneously is 3.75 watts. Maximum DC plate voltage is 450 V. A 12AU7A would
have worked fine: its maxima are 2.75 and 330. I may boose VDR+ to 250 V to give
the drivers a little extra oomph.

****     SMALL SIGNAL BIAS
SOLUTION       TEMPERATURE =   27.000
DEG C

NODE   VOLTAGE     NODE
VOLTAGE     NODE
VOLTAGE     NODE   VOLTAGE

(   1C)    9.0105  (
1G)    7.6727  (   1P)  235.3900
(   2C)  109.9600
(   2G)
108.6600  (   2P)  302.9800  (   3C)
-70.1920  (   3G)  -82.5550
(
3P)  179.6100  (   4C)  -70.1920  (
4G)  -82.5550  (   4P)  179.6100

(   5C)  -49.6790  (   5G)
-62.7190  (   6C)  -49.6930  (   6G)
-62.7340
(   7C)  -49.6930  (
7G)  -62.7340  (   8C)  -49.6930  (
8G)  -62.7340
(   9C)
1.1049  (   9S)  438.3000  (
10C)    1.1033  (  11C)    1.1033

(  12C)    1.1033  (  12S)
438.3000  (  A_6)    1.1080  (
A_7)    1.1080
(  BLK)
0.0000  (  BLU)    0.0000  (  BRN)
438.6700  (  B_6)  -62.7350
(  B_7)
-62.7350  (  B_8)  -62.7350  (
CRF)    1.1094  (  GRN)  438.6700

(  INP)    0.0000  (  ORN)
-82.5560  (  VA6)    6.6810  (  VB2)
-102.4900
(  VB3)  435.7900  (  VB4)
367.8200  (  VB6)   10.5180  (  VBB)
-82.5560
(  VBS) -115.0000  (  VD-)
-305.0000  (  VD1)  208.8900  (  VIO)
438.6700
(  VLT)  -82.5560  (
VOP)   12.5000  (  YEL)  438.6700  (
SPKR)    0.0000
( VBIN)  446.0000  (
VDR+)  204.9200  ( VDR-) -257.3600  (V+420)  413.8200

(V+450)  438.6700  (BIAS_5) -62.7190 (BIAS_6)  -62.7350
(BIAS_7)  -62.7350
(BIAS_8) -62.7350 ($N_0001)
7.6727
($N_0002)    7.6723

Norman Koren Vacuum tube
audio page | Photography
page
Feedback
and Fidelity | Improved
vacuum tube models for SPICE simulations

Images and text copyright © 2000-2015 by Norman
Koren.
Norman Koren lives in Boulder, Colorado, founded
Imatest LLC in 2004, previously worked on magnetic recording technology.
He has been involved with photography since 1964.